Distortion tolerant linear phase modulations

ABSTRACT

A signal and information transmission system for communications or radar and a method of achieving distortionless transmission of linear phase shift keying signals amplified by nonlinear power amplifiers or distorted by up-converters and RF circuits or the channel. Error correcting codes are used to jointly overcome both the distortion by the power amplifiers or up-converters and RF circuits or the channel and the noise. The modulator generates properly pulse shaped PSK signal, which does not have constant envelope. The signal-to-distortion power ratio is maximized to be above 20 dB at the fully saturated power amplifier output. The simplest linear receiver is provided to demodulate the received signal in the presence of radio distortion, channel distortion and noise. The decoder makes decision in the presence of radio distortion, channel distortion and noise. The method guarantees distortionless transmission of PSK signals for communications systems or radar employing power amplifiers of high DC-to-AC power conversion efficiency including class-F, class-E, class-D, class-C, or any of class-A, class-B, class-AB working in the saturation region. The method can achieve high bandwidth efficiency.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to generating a linearly modulated signal using phase shift keying, which can tolerate distortion in transmission, and achieving optimal or near optimal demodulation performance in the presence of distortion. The distortion can be caused by RF circuits including high power amplifiers and up-converters and mixers at radio frequency or by the channel. The invention may have applications in communications systems using phase shift keying, wireless communications, satellite communications, radar, and in particular in a communication terminal or radar which needs enough transmission power, or high DC-to-AC power conversion efficiency, or simple design for transmitter or receiver, or simple radio frequency circuit design.

2. Background and Description of Related Art

In recent years the data rate demanded for wireless communications has increased dramatically. Commercial products supporting 1 Gbps are already in service using millimeter wave and BPSK in ultra-broadband wireless communications in the last-mile. The data rate in wireless LAN is also increased significantly. The ultra-wideband (UWB) technology is under intense study for commercial applications or radar systems. It is desirable to create low complexity transceivers to support very high data rates or chip rates. The transmitter must have enough transmission power. For mobile terminals and transceivers operated on battery such as in wireless sensor networks, it is preferred to have high DC-to-AC power conversion efficiency. The high DC-to-AC power conversion efficiency can be achieved using the class-C, class-D, class-E or class-F power amplifiers. These power amplifiers cause non-negligible distortion to linear phase modulations and are traditionally not used in systems which employ non-constant envelope phase modulations such as phase shift keying. The transmitter or the receiver needs simple implementation including simple RF circuit to minimize the cost. The system needs high bandwidth efficiency and optimal or near optimal demodulation performance.

Constant envelope modulation methods have been widely used to tolerate distortion. This was achieved by keeping the envelope of the modulated signal as constant and embedding the information in the phase or the frequency of the modulated signal. A popular subset of constant envelope modulations is known as continuous phase modulation. The complexity in either the transmitter or the receiver is usually not low for constant envelope modulations. An equalizer is needed in the receiver to achieve optimal or near optimal demodulation performance. Because of the complexity, constant envelope modulations are not preferred when data rate or chip rate is high.

Herbst et. al. (as represented by U.S. Pat. No. 5,812,604) have invented a method to form constant envelope OQPSK signal using hard-limiting. The constant envelope OQPSK signal is easier to be up-converted to radio frequency.

Inspired by the results in the constant envelope OQPSK, which was independently obtained in 1996 by the inventor of this application, Jackson and Roos invented the implementation of the constant envelope OQPSK using MSK modulator. The implementation is represented by U.S. Pat. No. 6,301,310, which cannot give good power spectral density and causes large degradation in signal-to-noise power ratio for bit error performance.

In summary, previous methods have practiced generating constant envelope. They followed the traditional theory and practice that constant envelope signals can tolerate radio distortion.

SUMMARY OF THE INVENTION

This invention provides a novel method and system for signal and information, transmission which can overcome the distortion caused by RF circuits including power amplifiers and up-converters and mixers or by communication channels. Such power amplifiers can be in the class-C, class-D, class-E, class-F, or any of class-A and class-B and class-AB working in the saturation region. The invented method generates phase modulated signals in baseband and maximizes the signal-to-distortion power ratio at the output of the power amplifier or at the channel output. The maximization of the signal-to-distortion power ratio can be achieved using optimal filtering in base band. The invented filtering can be implemented using linear filter of finite impulse response or nonlinear filter. In the receiver, the invention uses a filter or filters matched to the base band filter or filters in the transmitter, which has the option to include the base-band equivalent model of the power amplifier or of the channel. The invention employs channel coding to overcome both the AWGN and the distortion caused by RF circuits such as power amplifiers and up-converters and mixers in the transmitter or by the channel. The implementation of the invention can have the simplest architecture for either the transmitter or the receiver. The invention can achieve high bandwidth efficiency. The invented method does not require linearity in the power amplifier or in the channel. The invention can achieve optimum or near optimum bit error performance. The invented method can significantly reduce the cost of communications terminals and increase the battery life by employing partially or fully saturated power amplifiers or power amplifiers of high DC-to-AC power conversion efficiency.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a wireless communications system in accordance with the invention;

FIG. 2 is a block diagram of the modulator for the invention;

FIG. 3 is a block diagram of the demodulator for the invention;

FIG. 4 is the power spectrum density for OQPSK using the invention with fully saturated power amplifier without filtering at the amplifier output to limit the out-of-band emission;

FIG. 5 is the bit error rate for OQPSK with hard decision employing the present invention;

FIG. 6 is the bit error rate for OQPSK with convolutional code or turbo code employing the present invention;

FIG. 7 is the bit error rate for $\frac{\pi}{4}$ QPSK employing the present invention with convolutional code or turbo code;

FIG. 8 is the bit error rate for QPSK employing the present invention with convolutional code or turbo code; and

FIG. 9 is the bit error rate for BPSK employing the present invention with convolutional code or turbo code.

DETAILED DESCRIPTION OF THE INVENTION

Referring to FIG. 1, a wireless communications system 10 in accordance with one embodiment of the present invention is illustrated. The system 10 includes a transmitter 12, a channel 14 and a receiver 16. The transmitter is in communication with the receiver in radio frequency (RF) through the channel.

The transmitter 12 has an information source 18, an encoder 20, a modulator 22, a high-power amplifier (HPA) with up-converter 24 and an antenna 26. The encoder performs encoding, which not only takes care of the additive noise in the channel but also overcomes the distortion. The distortion can be caused by either the high power amplifier and the up-converter with mixer 24 or the channel 14. The modulator 22 generates a modulated signal using phase shift keying. When the modulated signal is amplified and up-converted, the distortion at the output of the high power amplifier and up-converter 24 is minimized. The output signal of the high power amplifier and up-converter is fed into the antenna 26 and then transmitted at radio frequency. The modulated signal is transmitted through the channel 14.

The channel 14 can be a linear channel or a nonlinear channel. The channel adds noise to the signal. The channel can also cause distortion to the signal.

The receiver 16 includes an antenna 30, a down-converter 32, a demodulator 34, a decoder 36 and an information sink 38. The antenna receives the radio frequency signal with noise and distortion. The down-converter removes the carrier and gives the baseband signal with noise as the output. The demodulator performs packet detection, synchronization and filtering or correlation and recovers the signal in the presence of noise and distortion. The decoder makes decision for the information bits. The decoded bits are passed to the information sink.

The high power amplifier 24 or the channel 14 can cause severe distortion to the signal. For example, the power amplifier can be any of class-C, class-D, class-E and class-F. The power amplifier can be fully saturated and can completely remove the amplitude variation. In the traditional PSK modulations the amplitude variation is essential and the power amplifier is required to have good linearity for distortionless transmission. Consequently, traditional communications systems using PSK are well known for not being able to employ power amplifiers working in moderately or deeply saturated region. However, saturated power amplifiers can achieve much higher DC-to-AC power conversion efficiency. The high DC-to-AC power conversion efficiency is ideal in mobile communications, ultra-broadband wireless and satellite communications, and sensor networks.

The encoder 20 adds redundancy to the information sequence. The redundancy can help to overcome both the noise and the distortion introduced by either the power amplifier and RF circuits or the channel. The encoding methods are preferred to be, but not limited to, the following: (a) Turbo codes; (b) Convolutional codes; (c) Block codes; (d) Low density parity check codes.

The modulator 22 converts the discrete-time sequence at the encoder output to a continuous time waveform. The structure of the modulator is shown in FIG. 2. The map circuit 42 takes k=log₂ M bits at a time and maps these bits into the phase value using the Gray coding scheme. The phase value is applied to both the inphase channel and the quadrature channel. The quadrature channel signal can be delayed by half a symbol time by the delay module 44 when OQPSK is employed. Finite impulse response (FIR) filters 46 and 48 are employed in both channels for pulse shaping. The pulse shaping is to increase the spectral efficiency at the high power amplifier output even in the presence of strong distortion caused by the high power amplifier.

The coefficients of the FIR must be so carefully chosen that after the modulated waveform is amplified by the power amplifier, the distortion measured at the power amplifier output is minimized. There are many shaping functions which can satisfy this requirement. We recommend the square root raised cosine function with the duration as Lε[2,6] symbols and the roll-off factor βε[0.9,1.0]. Window functions in signal processing by Hanning, Hamming and Blackman can also be used as the shaping pulse. Let the pulse shaping function be p(t). Let the impulse response function of the high power amplifier be p₁(t). If p₁(t) is unknown to the designer, the shaping pulse p(t) should be chosen so that the Nyquist criterion can be satisfied, i.e., $\begin{matrix} {{x\left( {t = {kT}_{s}} \right)} = \left\{ \begin{matrix} 1 & {{{{If}\quad k} = L};} \\ 0 & {Else} \end{matrix} \right.} & (1) \end{matrix}$ where x(t) is the convolution of p(t) and p(LT_(s)−t). If p₁(t) is known to the designer, the Nyquist criterion should be satisfied with x(t) as the convolution of g(t)=p(t)*p₁(t) and g(T−t), where T is the duration of g(t).

The inphase signal is multiplied by the module 50 for the IF carrier cos(ωt). The quadrature signal is multiplied by the module 52 for the IF carrier sin(ωt). The signals in the inphase channel and the quadrature channel are added together by the module 54, and fed into the high power amplifier 24. The input signal to the high power amplifier 24 can be written as $\begin{matrix} \begin{matrix} {{s(t)} = {{\sum\limits_{k}{{p\left( {t - {kT}_{s}} \right)}\quad{\cos\left( \phi_{k} \right)}\quad{\cos\left( {2\quad\pi\quad{ft}} \right)}}} -}} \\ {\sum\limits_{k}{{p\left( {t - {kT}_{s} - \tau} \right)}\quad{\sin\left( \phi_{k} \right)}\quad{\sin\left( {2\quad\pi\quad{ft}} \right)}}} \end{matrix} & (2) \end{matrix}$

-   -   where {φ_(k)} is the phase sequence obtained using the Gray         encoding.

The power amplifier amplifies the signal to the desired power level and performs the up-conversion of the signal to the carrier frequency. The power amplifiers can be in the class-C, class-D, class-E or class-F. The power amplifier can also be any of the class-A, class-B, or class-AB working in the saturation region to gain high DC-to-AC power conversion efficiency.

Table 1 lists the signal-to-distortion power ratio measured at the fully saturated power amplifier output. The input to the fully saturated power amplifier is the modulated signal using the modulator in FIG. 2 for OQPSK. The square root raised cosine function with the roll-off factor β, among many good functions, serves as the pulse shaping function. It can be seen that the distortion to the signal at the fully saturated power amplifier output is minimized to be negligible. The distortion can be minimized to be negligible for the system in FIG. 1 employing the modulator in FIG. 2 using M -ary PSK, including BPSK, QPSK, OQPSK, π/4 QPSK, 8-PSK and 16-PSK.

The structure of the demodulator is shown in FIG. 3. The down-converter output is sampled by the module 62 at f, ≧4R_(s) samples per second, where R_(s) is the symbol rate. The signal in the in-phase channel is filtered by the filter 64, which is matched to the pulse shaping filter. The signal in the quadrature channel is filtered by the filter 66, which is also matched to the pulse shaping filter. The correlator 68 and the correlator 70 correlate the received signal with the preamble in each packet for packet detection and parameter estimation. The packet detector 72 detects each packet. The parameter estimator 74 estimates the symbol time, the carrier phase and the frequency offset. The tracking loop 76 is initialized using the symbol time estimate, the carrier phase estimate and the frequency offset estimate. At the tracking loop output, the errors are minimized for symbol time, carrier phase and frequency offset. TABLE 1 Signal-to-distortion power ratio of OQPSK signal amplified by fully saturated power amplifier. The shaping pulse is the square root raised cosine function with the roll-off factor β and the duration of L symbols. β 1.0 0.9 0.8 0.7 0.6 0.5 0.4 0.3 L = 8 21.38 19.86 18.43 17.11 15.92 14.84 13.88 13.03 L = 6 21.33 19.81 18.37 17.06 15.89 14.83 13.87 13.05 L = 4 21.34 19.84 18.40 17.06 15.85 14.80 13.90 13.17

The demodulator output is fed into the decoder 78. The decoder can be an iterative decoder for the turbo code, or the Viterbi decoder for the convolutional code, or the decoder for the low density parity check code, or the decoder for the block code.

FIG. 4 shows the power spectrum density in the solid line for a communications system or radar system employing the present invention with a fully saturated power amplifier. There is no band-limiting filter either in the power amplifier or after it in obtaining the power spectrum density. It can be seen that the sidelobe is below −26 dB. The sidelobe can be reduced by adding a band-limiting filter at the power amplifier output, when out of band emission needs further reduction of the sidelobe. For comparison, the power spectrum densities are also plotted in the dash-dot line for the traditional OQPSK using the square shaping pulse and a linear power amplifier, and in the dot line for MSK.

FIG. 5 shows the uncoded bit error rate for the communications system in FIG. 1. The modulation scheme is the OQPSK and the high power amplifier is fully saturated, which can be the class-C, class-D, class-E or class F power amplifiers, or any of the class-A, class-B, class-AB power amplifiers working in the saturated region. It can be seen that at BER=10⁻⁵, our method minimizes the SNR degradation introduced by the 1 fully saturated power amplifier to 0.1 dB. When the square root raised cosine function is employed as the pulse shaping function, the roll off factor is preferred to be close to 1.

FIG. 6 shows the coded bit error rate for the communications system in FIG. 1 employing OQPSK and convolutional code or turbo code. The power amplifier is fully saturated. For comparison, the dash line and the solid line show the performance for the ideal linear power amplifier. The rate is ½ and the constraint length is K=7 for the convolutional code. The Viterbi decoder is employed. The performance is better when the constraint length increases. The rate is ½ and the constraint length is K=4 for the turbo code. Three iterations are employed in the decoder. It can be seen that the degradation of the signal-to-noise power ratio caused by the fully saturated power amplifiers is negligible when the square root raised cosine filter is employed. The turbo code outperforms the convolutional code. When the pulse shaping filter in the IS-95 is employed, the fully saturated power amplifier causes the SNR to degrade by 0.5 dB at BER=10⁻⁵ for the convolutional coded system, or by 0.7 dB at BER=10⁻⁶ for the turbo coded system.

FIG. 7 is the bit error rate for $\frac{\pi}{4}$ QPSK employing the present invention with convolutional code or turbo code. The rate is ½ and the constraint length is K=7 for the convolutional code. The Viterbi decoder is employed. The performance is better when the constraint length increases. The rate is ½ and the constraint length is K=4 for the turbo code. Three iterations are employed in the decoder. It can be seen that the degradation of the signal-to-noise power ratio caused by the fully saturated power amplifiers is negligible.

FIG. 8 is the bit error rate for QPSK employing the present invention with convolutional code or turbo code. The rate is ½ and the constraint length is K=7 for the convolutional code. The Viterbi decoder is employed. The performance is better when the constraint length increases. The rate is ½ and the constraint length is K=4 for the turbo code. Three iterations are employed in the decoder.

FIG. 9 is the bit error rate for BPSK employing the present invention with convolutional code or turbo code. The rate is ½ and the constraint length is K=7 for the convolutional code. The Viterbi decoder is employed. The performance is better when the constraint length increases. The rate is ½ and the constraint length is K=4 for the turbo code. Three iterations are employed in the decoder. 

1. A signal and information transmission system and a method for communications or radar overcoming the distortion that can be caused by RF circuits including any of power amplifiers or up-converters or mixers or by the channel, the system comprising: means for using error correcting codes to combat both the distortion caused by either RF circuits including power amplifiers and up-converters and mixers or the channel and the noise at the same time; means for constructing phase modulated signals which can tolerate the distortion caused by the saturation of power amplifiers or the distortion caused by RF circuits such as up-converters and mixers or the channel; means for minimizing the distortion caused by saturated power amplifiers and RF circuits to phase-shift-keyed signals at the output of one saturated power amplifier or many saturated power amplifiers or RF circuit; means for employing any of class-C, class-D, class-E, or class-F power amplifiers, or class-A, class-B, class-AB power amplifiers working in the saturated region for signal power amplification or transmission; means for simplifying the processes of design, development and manufacturing of power amplifiers, up-converters, mixers, RF circuits, radio terminals, communications terminals and radar; means for reducing the radio cost, power amplifier cost, up-converter cost, mixer cost and terminal cost; means for increasing the reliability of power amplifiers and RF circuits and the reliability of radio terminals; means for achieving high spectral efficiency in the presence of distortion caused by power amplifiers, up-converters, mixers and RF circuits; means for simplifying the terminal design, development and manufacturing; means for increasing the battery life for communication terminals operated on battery; means for filtering the received signal with distortion and noise by filters that are matched to the filters in the transmitter; means for detecting the received signal; means for estimating parameters in the received signal; means for tracking the received signal in the presence of distortion and noise; means for demodulating the received signal using the error correcting codes; and means for achieving the bit error rate performance which is almost the same as the ideal linear BPSK bit error rate when any of class-F, class-E, class-D and class-C power amplifiers, or any of class-A, class-B, class-AB power amplifiers working in the saturated region is employed.
 2. A system as recited in claim 1, comprising the encoder for error correcting codes to add redundancy to the information bits by applying turbo code, convolutional code, block code, or low density parity check code to combat the distortion caused by power amplifiers and RF circuits or the channel and to overcome the noise in the channel.
 3. A system as recited in claim 1, comprising the modulator to take the input data stream and map the input data to the corresponding phase using the Gray coding for M-ary phase shift keying, M>=2, and variations of phase shift keying.
 4. A system as recited in claim 3, wherein said modulator constructs the modulated waveforms in orthogonal channels using finite impulse response filters.
 5. A system as recited in claim 4, wherein said modulator uses pulse shaping functions including but not limited to the square-root raised cosine function, raised cosine function, and functions of Hanning, Hamming and Blackman.
 6. A system as recited in claim 5, wherein said modulator constructs the linearly modulated waveforms of phase shift keying which do not have constant envelope in orthogonal channels.
 7. A system as recited in claim 6, wherein said modulator generates waveforms to minimize the distortion after being amplified by power amplifiers of class-F, class-E, class-D, class-C, or power amplifiers of class-A, class-B, class-AB working in the saturation region.
 8. A system as recited in claim 6, wherein said modulator constructs waveforms to minimize the distortion after being transmitted through the channel.
 9. A system as recited in claim 1, comprising up-converters and power amplifiers to convert the baseband signal to the signal centered at the carrier frequency and amplify the signal.
 10. A system as recited in claim 9, wherein said power amplifiers are class-F, class-E, class-D, class-C, or class-A, class-B, class-AB working in the saturation region, for high DC-to-AC power conversion efficiency.
 11. A system as recited in claim 1, comprising an antenna or multiple antenna for transmission and reception of the signal.
 12. A system as recited in claim 1, comprising a down-converter to convert the received signal to baseband.
 13. A system as recited in claim 1, comprising the sampling circuit to sample the received signal.
 14. A system as recited in claim 1, comprising filters in the receiver which are matched to the shaping filters in the transmitter or to the convolution of the impulses response functions of the shaping filter and the power amplifiers and the RF circuits in the transmitter.
 15. A system as recited in claim 1, comprising correlators to correlate the received signal in orthogonal channels with the prestored waveforms or signals.
 16. A system as recited in claim 1, comprising a signal detector which uses the output signals of the correlators to detect the signal or packet.
 17. A system as recited in claim 1, comprising parameter estimators to estimate the carrier frequency, phase and symbol timing, using the output of correlators.
 18. A system as recited in claim 1, comprising a tracking loop which uses the estimated carrier frequency, phase and symbol timing in the initialization of the loop to track the signal.
 19. A system as recited in claim 1, comprising a decoder which can be the iterative decoder, the maximum likelihood decoder or the block decoder.
 20. A system as recited in claim 19, wherein said decoder overcomes the distortion caused by power amplifiers or up-converters or RF circuits in the transmitter or by the channel and combats the noise in the channel, the decoding algorithm is optimal to the noise distribution.
 21. A system as recited in claim 1, comprising a decoder which can achieve optimal or near optimal bit error rate performance in the presence of both distortion caused by RF circuits including the power amplifiers and the up-converters and mixers or by the channel, and noise such as AWGN.
 22. A method of minimizing the distortion caused by at least one saturated power amplifier or up-converter or RF circuit to phase modulated signals, the method comprising of the steps of: mapping k bits d=(d₁,d₂, . . . ,d_(k)) at a time to one of M=2^(k) phases, the mapped phase is φ(d), which is a symbol; feeding the coefficient cos φ(d) to a first finite impulse response filter for summing a series of the delayed filter coefficients {p(j)} weighted by the symbol coefficients {cos φ(d)} to form the filtered in-phase signal; and feeding the coefficient sin φ(d) to a second finite impulse response filter for summing a series of the delayed filter coefficients {p(j)} weighted by the symbol coefficients {sin φ(d)} to form the filtered quadrature signal.
 23. Two pulse shaping filters as recited in claim 22, the first pulse shaping filter to filter the in-phase signal and the second pulse shaping filter to filter the quadrature signal, with the impulse response function p(t) of L≧2 symbols long, and the pulse shaping function is preferred to satisfy: ${x\left( {t = {kT}_{s}} \right)} = \left\{ \begin{matrix} 1 & {{{{If}\quad k} = L};} \\ 0 & {Else} \end{matrix} \right.$ where x(t) is the convolution of p(t) and p(LT_(s)−t), T, is the symbol time; when the impulse response p, (t) of the saturated radio or the channel is known to the designer, the function x(t) is the convolution of g(t)=p(t)*p₁(t) and g(T−t), where T is the duration of g(t).
 24. Two pulse shaping filters as recited in claim 22, the impulse response function of each filter is any of the square-root raised cosine function with the roll-off factor β close to 1.0, the window functions of Hanning, Hamming, and Blackman, or any function satisfying the condition in claim
 23. 25. A modulator implementing phase shift keying as recited in claim 22, the envelope of the modulated signal is not constant.
 26. A method of achieving high spectral efficiency through proper pulse shaping in the presence of distortion caused by RF circuits including any of saturated power amplifier or up-converter or mixer or RF circuit for communications systems or radar employing phase shift keying.
 27. A linear demodulator to demodulate the received signal in the presence of both distortion and noise, the linear demodulator comprising: a first down-converter to convert the received signal in the in-phase channel to baseband; a second down-converter to convert the received signal in the quadrature channel to baseband; a sampling circuit to sample the baseband signals in both the in-phase channel and the quadrature channel; an analog-to-digital converter to convert the sampled baseband signals to digital signals; a first finite impulse response filter matched to the shaping filter for the in-phase channel in the transmitter, the filter performs low-pass filtering for the samples in the in-phase channel; a second finite impulse response filter matched to the shaping filter for the quadrature channel in the transmitter, the filter performs low-pass filtering for the samples in the quadrature channel; a signal detector to detect the incoming signal using the matched filter output; a parameter estimator to estimate the symbol timing, carrier frequency offset and carrier phase using the matched filter output; a tracking loop to track the carrier frequency, carrier phase and the symbol timing; and a decoder to make decisions on the received symbols using the tracking loop output.
 28. A method of simplifying the design, development, manufacturing and testing of power amplifiers, up-converters, mixers, RF circuits, radio terminals and communications terminals for communications systems or radar employing phase shift keying.
 29. A method of increasing the reliability of power amplifiers, RF circuits, radio terminals and communications terminals for communications systems or radar employing phase shift keying.
 30. A method of reducing the power amplifier cost, up-converter cost, radio cost and terminal cost for communications systems or radar employing phase shift keying.
 31. A method of increasing the battery life for communications terminals operated on battery.
 32. A method of reducing the power consumption in transmitters for communications systems or radar employing phase shift keying. 